Driver circuit, corresponding device, apparatus and method

ABSTRACT

A (pre) driver circuit includes first and second output terminals configured to be coupled to a power transistor. A differential stage has non-inverting and inverting inputs for receiving an input voltage. The input voltage is replicated as an output voltage across the first and second output terminals as a drive signal for the power transistor. The differential stage includes a differential transconductance amplifier in a voltage follower arrangement configured to provide continuous regulation of a voltage at the first output terminal with respect to the second output terminal.

CROSS-REFERENCE TO RELATED APPLICATIONS

This application claims priority to Italian Patent Application No.102016000119626, filed on Nov. 25, 2016, which application is herebyincorporated herein by reference.

TECHNICAL FIELD

The description relates to driver circuits as well as a correspondingdevice, apparatus and method.

BACKGROUND

Pre-driver stages are extensively used in motor control ICs using ahalf/full-bridge topology; in such arrangements, power FETs are drivenby causing large currents to flow through them in order to energize e.g.external coils such as motor windings.

Integrated circuits (ICs) with embedded pre-driver stages can bespecifically designed to enable electro-mechanic (“mechatronics”) motorcontrol solutions for a range of motor control applications.

In such applications, shaping appropriately the motor phase voltagesapplied to power FET control is a point of interest. For instance, abasic issue may arise in controlling effectively a complex capacitiveload as an external FET, e.g., by shaping its gate-to-source voltage.

A conventional approach may involve a constant source/sink currentdriver configuration operating in an open loop fashion.

In view of the continuous activity in that area, as witnessed, e.g., bydocuments such as US2015/0349772 A1, a demand is still felt for improved(pre)driver arrangements.

SUMMARY

One or more embodiments may be used in various applications such as,e.g., electric motor control. One or more embodiments contribute tomeeting the demand for improved (pre)driver arrangements.

According to one or more embodiments, improved (pre)driver arrangementsare achieved by a circuit having the features set forth in the claimsthat follow.

One or more embodiments may also relate to a corresponding device (e.g.an IC driver for an electric motor), corresponding apparatus (e.g. anelectric motor equipped with a device according to one or moreembodiments) and a corresponding method.

One or more embodiments may offer one or more of the followingadvantages: an operational transconductance amplifier (OTA) topologythat facilitates achieving gate voltage shaping from a stabilitystandpoint; facilitated integration of a standby mode configurationusing the same output mirrors used for regulation; a compact structure,with area consumption depending primarily on power stages used to drivean external load (e.g. FET), namely the current capability required bythe application; continuously driving the gate voltage with a voltagecontrolled current source (e.g. an OTA) allows output voltage shapingusing a simple input structure that can be eventually trimmed to improveregulation accuracy.

BRIEF DESCRIPTION OF THE DRAWINGS

One or more embodiments will now be described, by way of example only,with reference to the annexed figures, wherein:

FIG. 1 is a general block diagram of one or more embodiments;

FIG. 2 is a detailed block diagram of one or more embodiments;

FIG. 3 is a detailed block diagram showing possible features of one ormore embodiments; and

FIG. 4 is a time diagram exemplary of possible operation of one or moreembodiments.

DETAILED DESCRIPTION OF ILLUSTRATIVE EMBODIMENTS

In the ensuing description, one or more specific details areillustrated, aimed at providing an in-depth understanding of examples ofembodiments. The embodiments may be obtained without one or more of thespecific details, or with other methods, components, materials, etc. Inother cases, known structures, materials, or operations are notillustrated or described in detail so that certain aspects ofembodiments will not be obscured.

Reference to “an embodiment” or “one embodiment” in the framework of thepresent description is intended to indicate that a particularconfiguration, structure, or characteristic described in relation to theembodiment is comprised in at least one embodiment. Hence, phrases suchas “in an embodiment” or “in one embodiment” that may be present in oneor more points of the present description do not necessarily refer toone and the same embodiment. Moreover, particular conformations,structures, or characteristics may be combined in any adequate way inone or more embodiments.

The references used herein are provided merely for convenience and hencedo not define the extent of protection or the scope of the embodiments.

The block diagram of FIG. 1 is exemplary of a (e.g., integrated—IC)pre-driver (e.g., integrated) circuit 100 having output terminals Gxyand Sxy for coupling to the gate and source terminals of an external(power) FET T for driving a load L possibly including e.g. inductivecomponents L1, L2 and resistive components R.

A circuit according to one or more embodiments may be used e.g. fordriving the high-side (HS) and low-side (LS) FETs which control aterminal such as one of the “phases” of an electric motor such as abrushless motor. The inductance L2 and the resistance R in FIG. 1 maythus be representative of such a motor phase, possibly having one ormore external R-C “snubber” networks associated therewith.

Operation of a (pre)driver arrangement as exemplified in FIG. 1 mayinvolve producing between two terminals Gxy and Sxy intended to becoupled e.g. to the gate and source terminals of the FET T a signal Vgsthat “copies” an input signal Vin (e.g. a PWM-modulated signal) appliedacross the inputs (non-inverting and inverting) of a differential stage10, e.g. an operational transconductance amplifier (hereinafter,briefly, OTA).

In one or more embodiments, the exemplary arrangement of FIG. 1 mayinclude input current source/sink generators SR+, SR− coupled to an e.g.high-impedance (non-inverting) input of the stage 10 in order to act aspositive and negative slew rate control, as discussed in the following.

In one or more embodiments, the high-impedance input (+) of the stage 10may not affect such a slew rate control because its input current isnegligible with respect to that of source/sink generators SR+, SR−.

Also, the input (−) of the stage 10 may connect to the output terminalof the stage 10, so that the current at such input (−) can besubstantially higher than the current at input (+).

This condition is otherwise optional and not mandatory.

The exemplary arrangement of FIG. 1 may be powered by a (voltage) supplyCP and include an input capacitor C10 across the inputs of thedifferential stage 10.

One or more embodiments may involve an operational transconductanceamplifier OTA arrangement based on a “voltage follower” approach inorder to continuously regulate the voltage at terminal Gxy (e.g. thegate voltage of the FET T) with respect to the terminal Sxy in order toachieve better and smoother voltage control during switching transients.

An OTA regulation scheme may provide the benefit of beingstability-compensated by a capacitance (e.g. the gate-source capacitanceCgs of the external FET T) acting as dominant pole for the wholestructure: this facilitates implementing a loop regulation in unitaryfeedback configuration. In one or more embodiments, stability andtransient response of an arrangement as exemplified in FIG. 1 may beobserved to be dependent on the capacitive component of the FET T asseen from the circuit 100 e.g. on Cgs plus Cgd(Vgd), where Cgd(Vgd)denotes the gate-drain capacitance, in turn a function of the gate-drainvoltage Vgd.

Under these conditions, while the “external” FET T may be a distinctelement from the circuit of one or more embodiments, the dominant polefor the whole structure may be due to the external capacitanceassociated therewith.

Consequently, in one or more embodiments, a circuit architecture with acertain gain-bandwidth product (GBWP) may be able to “shape” thegate-to-source voltage Vgs starting from an internal voltage (Vin inFIG. 1) that can be easily driven using internal current source/sinkgenerators e.g. the slew rate source SR+ and slew rate sink SR−generators as depicted in FIG. 1.

Possible exemplary implementations of the general layout of FIG. 1 arepresented in FIGS. 2 and 3. In FIG. 2, the capacitive component of theFET T is explicitly shown and designated Cload.

Also, the various MOSFETs illustrated in FIGS. 2 and 3 may include arepresentation of their associated body diodes while the various currentgenerators illustrated may be implemented according to well knownprinciples in the art.

In one or more embodiments as exemplified in FIGS. 2 and 3, in the placeof a conventional OTA arrangement with two high impedance inputterminals, a floating gate driver may be implemented with only onesingle high impedance input branch, with the input signal Vin applied tothe gate of a MOSFET M7, e.g. across the input capacitor C10.

In one or more embodiments, the MOSFET M7 may be coupled to the outputterminal Gxy, e.g. via the current path (source-drain) of a furtherMOSFET M8.

The negative input branch of the embodiments as exemplified in FIGS. 2and 3 corresponds to the drain terminal of MOSFET M8, which connects tothe output terminal Gxy of the OTA. In contrast to a conventional OTAarrangement, the negative input does not need to have necessarily highimpedance, because the OTA output has sufficient capability of supplying(e.g. via the current path through M8) an amount of current at the OTAnegative input without showing malfunctions.

A diode D4 coupled between the source (coupled to the diode anode) andthe gate (coupled to the diode cathode) of M7, may be provided with thepurpose of protecting the gate oxide of MOSFETS M7 and M8.

For instance (as further discussed with reference to FIG. 4), afterhaving powered up M8 and M7, DC regulation may be set in order to haveacross the output terminals Gxy, Sxy a voltageVgs=Vin−Vgs_M7−Rdson_M8·I2, where: Vgs_M7 is the gate-source voltage atM7, Rdson_M8 and I2 are the Rdson resistance and the current through M8.

Due to the selected OTA topology, at each Vgs operating point a DCquiescent current IQ will flow through the current paths (e.g. thesource-drain paths) of two MOSFETs M11 and M10 acting as high side andlow side (current) mirrors, respectively, having their current pathsarranged in series and with the terminal Gxy between M11 and M10, withthe source-to-drain drop across M10 corresponding to the gate-sourcevoltage Vgs of the external FET T.

The gain stage may be configured in order to achieve a good trade-offbetween internal current consumption (IQ) and transient response intracking variations in the input signal.

One or more embodiments as exemplified in FIG. 2 may provide for the Vgsvoltage between Gxy and Sxy being finally clamped at a maximum value.

To that effect, an embedded DC voltage clamp may be provided e.g. as aset of Zener diodes D1, D2 and D3 arranged in series with their anodesfacing towards Sxy and their cathodes towards a MOSFET M1 having itsgate set between the Zener diodes D1, D2 and D3 and a current generatorI1 coupled to the supply line CP. The MOSFET M1 is arranged with itscurrent path (source—drain) between the supply line CP and the cascadedarrangement of the mirrors M11 and M10).

The clamping effect will thus result in the Vgs voltage being clamped ata value: Vgs_max=Vz_D1+Vz_D2+Vz_D3−Vgs_M1, where: Vz_Di i=1, 2, 3 arethe Zener voltages of the diodes D1, D2 and D3, and Vgs_M1 is thegate-source voltage of M1.

During the switch-on and -off transients, the gate-to-source voltage ofT may be sensed in order to generate an error current signal Ierr+,Ierr− indicative (e.g. proportional) to the error between the input Vinand the output Vgs as in a unity gain OTA buffer configuration.

An exemplary representation of a corresponding AC signal path is shownin FIG. 2 by assuming a small signal variation in Vin.

In one or more embodiments, a corresponding arrangement may include amirror stage including a MOSFET M4 (mirroring M11) and a MOSFET M9(mirroring M10), with a further MOSFET M5 having its current path(source drain path) set between the current paths of M4 and M9.

In the exemplary representation of FIG. 2, the mirror MOSFET M4 may bereferred to a line VL extending between M1 and M11, with two MOSFETs M2and M3 arranged with their current paths in series between VL and thegate of M5, with a current generator I3 active on a line connecting thegate of M5 to the line VL via a (bias) voltage source V_(B).

In one or more embodiments, a current generator I2 may be active on aline between VL and (the drain of) M7 with the error current Ierr+,Ierr− flowing (in both directions) over a line linking an intermediatepoint between M4 and M5 and an intermediate point between I2 and M7.

The MOSFET M7 will thus be in a position to unbalance (e.g. via M8) thetwo output mirrors (M4-M11 and M9-M10) in order to provide a current tocharge/discharge the external FET T.

In one or more embodiments, the input slew rate generators SR+ (currentI4) and SR− (current I5) may act between the line VL and the gate of M7(to which the input voltage Vin is applied) and between the gate of M7and Sxy, respectively.

This loop strategy facilitates reducing the transistor (e.g. MOSFET)lengths since negative feedback can easily recover internal currentmirror offset.

Therefore, in one or more embodiments, area consumption may be mainlydue to the final mirror stage (M4-M11 and M9-M10), which can be tailoredby taking into account current requirements for various applications.

By combining this input flexibility with an OTA topology, a compactstructure with a gate voltage shaping feature may be achieved, which canbe tailored (e.g. as regards slew rate selection and time duration) inaccordance with the application requirement.

The block diagram of FIG. 3 is exemplary of one or more embodiments of agate driver arrangement sharing the same basic layout of FIG. 2.

For that reasons, parts or elements corresponding to parts or elementsalready described in relation with the previous figures are indicated inFIG. 3 with the same references appearing in the previous figureswithout repeating a corresponding description.

It will otherwise be appreciated that various features distinguishingthe exemplary arrangement of FIG. 3 over the exemplary arrangement ofFIG. 2 may be applied individually, that is one independently of theothers: the combined discussion of these features provided in relationto FIG. 3 is thus for the sake of brevity only and is not to beconstrued in a limiting sense of the embodiments.

In comparison with FIG. 2, the gate driver arrangement of FIG. 3 (wherethe gate-drain and gate-sources capacitances Cgd and Cgs of the externalFET T are explicitly shown), includes possible variants such as e.g.substituting for one or more of the Zener diodes D1, D2, D3—e.g. for thediode D1—a cascaded arrangement of two bipolar transistors T1, T2 in adiode-like arrangement (with bases shorted to collectors) so that theVgs voltage may be clamped at a valueVgs_max=Vbe_T1+Vbe_T2+Vz_D2+Vz_D3−Vgs_M1, where Vbe_T1 and Vbe_T2 arethe base-emitter voltages of T1 and T2, respectively.

Also, in the gate driver arrangement of FIG. 3 (which may be suitablefor implementation with AlCu as top metal layer), the feedback loop maybe modified by providing a current generator I7 acting between theoutput drain of M8 and Sxy, while possibly protecting the Vgs of M7-M8with the diode D4.

In one or more embodiments, the current generators I2 and I7 may besubstantially identical and matched in order to have a better control onthe dominant pole as given by fp=K/(2π·Rpoly·Cgate), where thepolysilicon resistance Rpoly (explicitly shown in the figure) can be setin order to achieve a desired GBWP frequency behavior, K being thecurrent gain from OTA negative input at drain of M8 to the OTA output atGxy terminal.

The gate driver arrangement of FIG. 3 may include an OTA regulationblock 12 coupled to the mirrors M4-M11 and M9-M10 (and the line Vl andthe bias line of bias voltage B).

As in the case of the homologous circuitry of FIG. 2 including theMOSFETs M2, M3 and M5 and the associated components, the OTA regulationblock 12 of FIG. 3 may be stimulated by an input error current signalIerr+, Ierr− (in either direction) indicative e.g. proportional to thedifference between the “internal” Vin voltage and the “external” Vgsvoltage, so that the two output mirrors (M4-M11 and M9-M10) may beunbalanced (e.g. via M7 and M8) in order to provide a current tocharge/discharge the external FET without excessive cross conductionduring transients.

In one or more embodiments, a gate driver arrangement as exemplified inFIG. 3 may include a set of switches S1, S2 and S3, which may facilitatereducing internal current consumption after complete Vgs commutation.

In one or more embodiments the switches S1, S2 and S3 may be configuredto be selectively switched-off (that is made nonconductive) in order toselectively “isolate” the gates of M11 resp. M10 from: M4 resp. M9(switches S1); the negative side of bias voltage VB resp. Sxy (switchesS2); and the line VL resp. the gate of a MOSFET M17 (switches S3).

The MOSFET M17 is arranged with its current path coupled in seriesbetween a current generator I13 coupled to the line VL and the currentpath of a further MOSFET 18 set between the MOSFET M17 and Sxy.

In one or more embodiments, such an arrangement may permit to produce areference voltage Vgs18+Vgs17 (i.e. the sum of the Vgs voltages of M18and M17) for connection to the gate of M10 via a switch S3 as discussedin the following. This may be helpful when desiring that M10 may befully conductive as a result of the output voltage at Gxy havingconcluded its transition from VL to Sxy. Similarly, a voltage VBsubtracted from V1 may produce a reference voltage V1-VB for connectionof the gate of M11 via a switch S2. This may be helpful when desiringthat M11 may be fully conductive as a result of the output voltage atGxy having concluded its transition from Sxy to VL.

In one or more embodiments, a gate driver arrangement as exemplified inFIG. 3 may include a gate-off comparator 14 configured for comparing theoutput signal on the output terminal Gxy and the signal at a pointbetween a current generator I14 coupled to the line VL and a diodeconfiguration (e.g. MOSFET M16 with gate shorted to drain) arrangedbetween the gate-off comparator 14 and Sxy.

The gate-off comparator 14 may provide information concerning the OFFcondition of the external FET, e.g. in order to implement a dead timegeneration function aimed at avoiding High Side and Low Side FETcross-conduction on motor terminals.

As exemplified in the time diagram of FIG. 4, various phases may thusoccur during a complete switching period, including a STANDBY modefunctionality.

The time diagram of FIG. 4 is exemplary of a method of operating thecircuit of one or more embodiments where a switching period(corresponding e.g. to one cycle of a PWM-modulated input signal Vi) mayinvolve subsequent phases 200, 202, 204, 206, 208 including an ONregulation phase (ON PHASE 202) and an OFF regulation phase (OFF PHASE206) “interleaved” between standby phases, STDBY_L 200 and 208 andSTDBY_H 204.

In one or more embodiments STDBY_H at 204 (with S1 off, S2 on and S3off) may involve the output terminal Gxy pulled “up” while STDBY_L at200 and 208 (with S1 off, S2 off and S3 on) may involve Gxy pulled“down” (e.g. to Sxy).

In both instances that result may be achieved with low impedance and byusing the same power mirror employed for regulation.

Operation of the circuits exemplified herein (e.g. activating the slewrate generators SR+, SR−, turning the switches S1, S2 and S3 on and off,and so on) may be controlled by a logic circuit LC.

In one or more embodiments such a logic circuit may include e.g. alogical multiplexer configured to operate according to a coded signal(control code) devised to control the slew rate (positive or negative)at the input (high voltage) capacitance C10.

For instance, based on the threshold voltage Vth of the external FET andthe gm characteristics, the slew rate at the input capacitance C10 maycontrol the overall gate charge/discharge curve in order to smoothen theringing phenomenon during switching transitions using a real FET Rdsmodulation.

The diagram of FIG. 4 is exemplary of e.g. three slew rates applied withtime durations) e.g. t1, t2, t3 possibly programmable (e.g. via a SPIinterface) in order to get an optimal shaping.

The diagram of FIG. 4 is exemplary of possible operation where shapingthe gate charging/discharging profile during an ON PHASE e.g. mayinclude: an interval t1 from oV to Vth; an interval t2 close to theMiller plateau (this may facilitate damping the first RLC seriesresonance) an interval t3 from the Miller transition to full Vgs, whichis controlled to be not so fast in order to avoid a second criticalresonant behavior.

One or more embodiments may provide a circuit (e.g. 100), including:first (e.g. Gxy) and second (e.g. Sxy) output terminals coupleable to apower transistor (e.g. T). a differential stage (e.g. an OTA 10) havingnon-inverting and inverting inputs for receiving an input voltage (e.g.Vin) applied across the non-inverting and inverting inputs where theinput voltage is replicated as an output voltage (e.g. Vgs) across thefirst and second output terminals to provide a drive signal for thepower transistor, where the differential stage includes a differentialtransconductance amplifier (e.g. M7, M8) in a voltage followerarrangement providing continuous regulation of the voltage at the firstoutput terminal with respect to the second output terminal.

In one or more embodiments, the differential transconductance amplifierin a voltage follower arrangement may include at least one of: ahigh-impedance input (e.g. M7) having coupled therewith input currentsource resp. sink generators (e.g. SR+, SR− in FIG. 1; I4, I5 in FIGS. 2and 3) to provide slew rate control, and/or a further input (optionallya low-impedance input, a high-impedance input being also acceptable)coupled (e.g. via M8) to the first output terminal.

In one or more embodiments the differential transconductance amplifierin a voltage follower arrangement may include a first transistor (e.g.M7) having a control electrode (e.g. a gate in the case of FET, a basein the case of a bipolar transistor) to receive the input signal, thefirst transistor coupled to the first output terminal via the currentpath (e.g. source-drain in the case of FET, emitter-collector in thecase of a bipolar transistor) through a second transistor (e.g. M8).

One or more embodiments may include at least one of: a protection diode(e.g. D4) set between the control electrode of the first transistor andthe current path through the second transistor, and/or a currentgenerator (e.g. I7) between the output of the second transistor and thesecond output terminal.

In one or more embodiments the differential stage may include a voltageclamp (e.g. D1, D2, D3 FIG. 2; T1, T2, D2, D3 in FIG. 3) to clamp at apeak value the output voltage (e.g. Vgs) across the first and secondoutput terminals, the voltage clamp set between the second outputterminal and a supply line (e.g. CP) to the circuit.

One or more embodiments may include a sensor (e.g. M5 in FIG. 2; 12 inFIG. 3) configured for sensing the output voltage across the first andsecond output terminals and generating an error signal (for instanceIerr+, Ierr−) indicative of the error between the input voltage and theoutput voltage.

In one or more embodiments the differential stage may include high-side(e.g. M11) and low-side (e.g. M10) output current mirrors with the firstoutput terminal set there between.

One or more embodiments may include a switch set (e.g. S1, S2, S3)activatable (e.g. via LC) during high- and low-standby phases forselectively decoupling (e.g. switch S1) the output current mirrors fromthe differential transconductance amplifier while coupling (e.g. via theswitches S2, S3) the output current mirrors to a pull-up source (e.g. toVL, in a high-standby phase) and to a pull-down source (e.g. to Sxy, ina low-standby phase) for the first output terminal.

One or more embodiments may provide a driver device (e.g. a driver foran electric motor), including: a circuit according to one or moreembodiments, and a power transistor (e.g. T) having a control terminal(e.g. a gate in the case of FET, a base in the case of a bipolartransistor) and current path (e.g. source-drain in the case of FET,emitter-collector in the case of a bipolar transistor), the powertransistor having the control terminal and the current path coupled tothe first and second output terminals, respectively.

One or more embodiments may provide apparatus (e.g. electric motor),including: a driver device according to one or more embodiments, a load(e.g. a motor winding L) supplied via the power transistor.

One or more embodiments may provide a method of operating a circuitaccording to one or more embodiments, the method including (see e.g.200, 202, 204, 206, 208 in FIG. 4) varying the input voltage (and thusthe output voltage) through alternate on (e.g. 202) and off (e.g. 206)phases, the method including selectively (see e.g. t1, t2, t3 in FIG. 4)controlling (e.g. via LC) the slew rate (e.g. via SR+, SR−; I4, I5) atthe input of the differential stage during the alternate on and offphases.

Without prejudice to the underlying principles, the details andembodiments may vary, even significantly, with respect to what has beendisclosed by way of example only, without departing from the extent ofprotection.

The extent of protection is defined by the annexed claims.

What is claimed is:
 1. A circuit comprising: first and second outputterminals configured to be coupled to a power transistor; and adifferential stage having a non-inverting input and an inverting inputconfigured to receive an input voltage applied across the non-invertingand inverting inputs, wherein the input voltage is replicated as anoutput voltage across the first and second output terminals to provide adrive signal for the power transistor, wherein the differential stagecomprises a differential transconductance amplifier in a voltagefollower arrangement configured to provide continuous regulation of avoltage at the first output terminal with respect to the second outputterminal, wherein the differential transconductance amplifier in avoltage follower arrangement comprises a high-impedance input coupled tocurrent generators configured to provide slew rate control.
 2. Thecircuit of claim 1, wherein the high-impedance input is coupled a gateof a first transistor and to a gate of a second transistor, the secondtransistor coupled to the first output terminal.
 3. The circuit of claim1, wherein the differential transconductance amplifier in a voltagefollower arrangement comprises a first input coupled to the first outputterminal.
 4. A circuit comprising: first and second output terminalsconfigured to be coupled to a power transistor; and a differential stagehaving a non-inverting input and an inverting input configured toreceive an input voltage applied across the non-inverting and invertinginputs, wherein the input voltage is replicated as an output voltageacross the first and second output terminals to provide a drive signalfor the power transistor, wherein the differential stage comprises adifferential transconductance amplifier in a voltage followerarrangement configured to provide continuous regulation of a voltage atthe first output terminal with respect to the second output terminal,wherein the differential transconductance amplifier in a voltagefollower arrangement comprises a first transistor having a controlelectrode configured to receive an input signal, the first transistorcoupled to the first output terminal via a current path of a secondtransistor.
 5. The circuit of claim 4, further comprising a protectiondiode coupled between the control electrode of the first transistor andthe current path of the second transistor.
 6. The circuit of claim 4,further comprising a current generator coupled between an output of thesecond transistor and the second output terminal.
 7. The circuit ofclaim 1, wherein the differential stage comprises a voltage clampconfigured to clamp at a peak value the output voltage across the firstand second output terminals, the voltage clamp set between the secondoutput terminal and a supply line to the circuit.
 8. A circuitcomprising: first and second output terminals configured to be coupledto a power transistor; and a differential stage having a non-invertinginput and an inverting input configured to receive an input voltageapplied across the non-inverting and inverting inputs, wherein the inputvoltage is replicated as an output voltage across the first and secondoutput terminals to provide a drive signal for the power transistor,wherein the differential stage comprises a differential transconductanceamplifier in a voltage follower arrangement configured to providecontinuous regulation of a voltage at the first output terminal withrespect to the second output terminal, wherein the differential stagecomprises a voltage clamp configured to clamp at a peak value the outputvoltage across the first and second output terminals, the voltage clampset between the second output terminal and a supply line to the circuit,and wherein the voltage clamp comprises a first Zener diode coupledbetween the supply line and the second output terminal, wherein thevoltage clamp further comprises a first bipolar transistor coupled inseries with the first Zener diode.
 9. A circuit comprising: first andsecond output terminals configured to be coupled to a power transistor;a differential stage having a non-inverting input and an inverting inputconfigured to receive an input voltage applied across the non-invertingand inverting inputs, wherein the input voltage is replicated as anoutput voltage across the first and second output terminals to provide adrive signal for the power transistor, wherein the differential stagecomprises a differential transconductance amplifier in a voltagefollower arrangement configured to provide continuous regulation of avoltage at the first output terminal with respect to the second outputterminal; and a sensor configured to sense the output voltage across thefirst and second output terminals and to generate an error signalindicative of an error between the input voltage and the output voltage.10. A circuit comprising: first and second output terminals configuredto be coupled to a power transistor; and a differential stage having anon-inverting input and an inverting input configured to receive aninput voltage applied across the non-inverting and inverting inputs,wherein the input voltage is replicated as an output voltage across thefirst and second output terminals to provide a drive signal for thepower transistor, wherein the differential stage comprises adifferential transconductance amplifier in a voltage followerarrangement configured to provide continuous regulation of a voltage atthe first output terminal with respect to the second output terminal,wherein the differential stage comprises high-side and low-side outputcurrent mirrors, the first output terminal coupled between the high-sideoutput current mirror and the low-side current mirror.
 11. The circuitof claim 10, further comprising a switch set activatable duringhigh-standby and low-standby phases for selectively decoupling theoutput current mirrors from the differential transconductance amplifierwhile coupling the output current mirrors to a pull-up source and to apull-down source for the first output terminal.
 12. The circuit of claim11, wherein: the high-side output current mirror comprises a thirdtransistor and a fourth transistor; the low-side output current mirrorcomprises a fifth transistor and a sixth transistor; and the switch setcomprises a first switch coupled between a gate of the third transistorand a gate of the fourth transistor, a second switch coupled between agate of the fourth transistor and the pull-up source, a third switchcoupled between a gate of the fourth transistor and the pull-downsource, a fourth switch coupled between a gate of the fifth transistorand a gate of the sixth transistor, a fifth switch coupled between agate of the sixth transistor and the pull-up source, a sixth switchcoupled between a gate of the sixth transistor and the pull-down source.13. The circuit of claim 12, wherein: during the high-standby phase, thefirst, third, fourth and sixth switches are off and the second and fifthswitches are on; and during the low-standby phase, the first, second,fourth and fifth switches are off and the third and sixth switches areon.
 14. A system comprising: a power transistor; and a drivercomprising: a first output terminal coupled to a gate of the powertransistor, a second output terminal coupled to a first conductionterminal of the power transistor, a differential stage configured toreceive an input voltage across a first and second inputs of thedifferential stage, wherein the input voltage is replicated as an outputvoltage across the first and second output terminals, wherein thedifferential stage comprises a differential transconductance amplifierin a voltage follower arrangement configured to regulate a voltagebetween the first output terminal and the second output terminal,wherein the differential transconductance amplifier comprises ahigh-impedance input coupled to current generators configured to provideslew rate control.
 15. The system of claim 14, wherein the firstconduction terminal of the power transistor comprises a source terminal.16. A system comprising: a power transistor; a driver comprising: afirst output terminal coupled to a gate of the power transistor, asecond output terminal coupled to a first conduction terminal of thepower transistor, a differential stage configured to receive an inputvoltage across a first and second inputs of the differential stage,wherein the input voltage is replicated as an output voltage across thefirst and second output terminals, wherein the differential stagecomprises a differential transconductance amplifier in a voltagefollower arrangement configured to regulate a voltage between the firstoutput terminal and the second output terminal; and a load coupled tothe first conduction terminal of the power transistor.
 17. The system ofclaim 16, wherein the load comprises a brushless motor.
 18. A method ofoperating a driver coupled to a power transistor, the method comprising:receiving an input voltage across a non-inverting and inverting inputsof a differential stage having a transconductance amplifier in a voltagefollower arrangement, the input voltage alternating between on and offphases; producing a regulated output voltage between a gate of the powertransistor and a first conduction terminal of the power transistor basedon the input voltage with the transconductance amplifier; andselectively controlling a slew rate at the inputs of the differentialstage during the alternate on and off phases of the input voltage,wherein selectively controlling the slew rate comprises controlling: afirst current generator coupled between the non-inverting input of thedifferential stage and a voltage supply terminal, and a second currentgenerator coupled between the non-inverting input of the differentialstage and the power transistor.
 19. The method of claim 18, furthercomprising receiving an input signal at the non-inverting input of thedifferential stage with a control electrode of a first transistor andwith a control electrode of a second transistor, the first transistorcoupled to the gate of the power transistor via the second transistor.20. The method of claim 19, further comprising generating an errorsignal indicative of an error between the input voltage and theregulated output voltage at an error node coupled to the firsttransistor, the error node coupled to the gate of the power transistorvia a first current mirror, and to the first conduction terminal of thepower transistor via a third transistor, the third transistor and afourth transistor forming a second current mirror coupled to the gate ofthe power transistor.